Systems such as “Active phased-array antenna and antenna control unit” described in Japanese Published Patent Application No. 2000-236207 (hereinafter, referred to as Prior Art 1) have been suggested as examples of conventional phased-array antennas that employ a ferroelectric as a phase shifter.
Hereinafter, a conventional phased-array antenna will be described with reference to FIGS. 9 and 10.
Initially, operating principles of a conventional phase shifter are described with reference to FIGS. 9(a) and 9(b). FIGS. 9(a) and 9(b) are diagrams illustrating a phase shifter 700 that is suggested in the conventional phased-array antenna. FIG. 9(a) is a diagram illustrating a construction of the phase shifter 700, and FIG. 9(b) is a diagram showing permittivity changing characteristics of a ferroelectric material.
This phase shifter 700 includes a microstrip hybrid coupler 703 that employs a paraelectric material 701 as a base material, and a microstrip stub 704 that employs a ferroelectric material 702 as a base material and is formed adjacent to the microstrip hybrid coupler 703. This phase shifter 700 is constituted such that a phase shift amount of a high-frequency power that passes through the microstrip hybrid coupler 703 varies according to a DC control voltage which is applied to the microstrip stub 704.
In other words, the base material of the phase shifter 700 is composed of the paraelectric material 701 and the ferroelectric material 702. A rectangular loop-shaped conductor layer 703a is disposed on the paraelectric base material 701, and this loop-shaped conductor layer 703a and the paraelectric base material 701 form the microstrip hybrid coupler 703.
Further, two linear conductor layers 704a1 and 704a2 are disposed on the ferroelectric base material 702 so as to be located on extension lines of two opposed linear parts 703a1 and 703a2 of the rectangular loop-shaped conductor layer 703a and linked to one of the ends of the two linear parts 703a1 and 703a2, respectively. These two linear conductor layers 704a1 and 704a2 and the ferroelectric base material 702 form the microstrip stub 704.
Further, conductor layers 715a and 720a are disposed on the paraelectric base material 701 so as to be located on extension lines of the two linear parts 703a1 and 703a2 and linked to the other ends of the two linear parts 703a1 and 703a2, respectively.
This conductor layer 715a and the paraelectric base material 701 form an input line 715, and the conductor layer 720a and the paraelectric base material 701 form an output line 720.
Here, the one end and the other end of the linear part 703a1 on the loop-shaped conductor layer 703a are ports 2 and 1 of the microstrip hybrid coupler 703, respectively. On the other hand, the one end and the other end of the linear parts 703a2 of the loop-shaped conductor layer 703a are ports 3 and 4 of the microstrip hybrid coupler 703, respectively.
In the phase shifter 700 having the above-mentioned construction, when the DC control voltage is applied to the microstrip stub 704, the phase shift amount of the high-frequency power that passes therethrough varies.
Hereinafter, a detailed explanation of the phase shifter 700 will be given. In the phase shifter 700 having such a construction in which one reflection element (microstrip stub 704) is connected to the adjacent two ports (ports 2 and 3) of the properly-designed microstrip hybrid coupler 703, a high-frequency power that enters from the input port (port 1) is not outputted from the input port 1, but the high-frequency power upon which a power reflected from the reflection element has been reflected is outputted only from the output port (port 4). In the reflection from the microstrip stub 704 as the reflection element, a bias field 705 that is produced by the control voltage is in the same direction as that of a field produced by the high-frequency power that passes through the microstrip stub 704, as shown in FIG. 9(a). Therefore, as shown in FIG. 9(b), when the control voltage is changed, an effective permittivity of the microstrip stub 704 with respect to the high-frequency power varies adaptively. Accordingly, the equivalent electrical length of the microstrip stub 704 for the high-frequency power varies, and the phase on the microstrip stub 704 is changed.
In the case of common ferroelectric base materials, the bias voltage 705 that is required to change the effective permittivity of the microstrip stub 704 is in a rage of several kilovolts/millimeter to a dozen kilovolts/millimeter. Accordingly, a high frequency is not produced by the effective permittivity that is affected by a field formed by the high-frequency power which passes through the microstrip stub 704.
Next, a construction of the conventional phased-array antenna and its operating principles will be described with reference to FIGS. 10(a) and 10(b).
FIG. 10(a) is a diagram illustrating a construction of the conventional phased-array antenna 830, and FIG. 10(b) is a diagram showing directivities of the conventional phased-array antenna 830 in a case where a beam tilt voltage is applied and a case where the beam tilt voltage is not applied.
The conventional phased-array antenna 830 comprises plural antenna elements 806a-806d which are placed in a row at regular intervals on a dielectric base material, an antenna control unit 800, and a beam tilt voltage 820. The antenna control unit 800 comprises a feeding terminal 808 to which a high-frequency power is applied (hereinafter, referred to as an input terminal), a high frequency blocking element 809, and plural phase shifters 807a1-807a4.
In this conventional phased-array antenna 830, the antenna element 806a is connected to the input terminal 808, the antenna element 806b is connected to the input terminal 808 through one phase shifter 807a1, the antenna element 806c is connected to the input terminal 808 through two phase shifters 807a3 and 807a4, and the antenna element 806d is connected to the input terminal 808 through three phase shifters 807a2, 807a3, and 807a4, by means of a feeding line (hereinafter, referred to as a transmission line), respectively. The beam tilt voltage 820 is connected to the input terminal 808 through the high frequency blocking element 809.
It is assumed here that each construction of the phase shifters 807a1-807a4 is the same as that described with reference to FIG. 9, and the phase shifters 807a1-807a4 have the same characteristics.
In the phased-array antenna 830 having the above construction, the number of phase shifters 807 which are located between one of the antenna elements 806a-806d and the input terminal 808 is one larger than the number of phase shifters 807 which are located between the adjacent antenna element 806 and the input terminal 808, respectively, and further, all of the phase shifters 807 have the same characteristics. Therefore, as shown in FIG. 10(b), the control of the antenna's directivity (beam tilt) is performed by one beam tilt voltage 820.
The control of the antenna directivity will be described in more detail. For example, assuming that each of the phase shifters 807a1-807a4 delays the phase of the high-frequency power that passes through each phase shifter by a phase shift amount Φ and the adjacent phase shifters 807 are spaced by a distance d, respectively, the high-frequency power that has entered the antenna element 806a is supplied to the input terminal 808 with no phase change, as shown in FIG. 10(a). In contrast to this, the high-frequency power that has entered the antenna element 806b is supplied to the input terminal 808, with its phase being delayed by the phase shifter 807a1 by a phase shift amount Φ. The high-frequency power that has entered the antenna element 806c is supplied to the input terminal 808, with its phase being delayed by the phase shifters 807a3 and 807a4, by a phase shift amount 2Φ. Further, the high-frequency power that has entered the antenna element 806d is supplied to the input terminal 808, with its phase being delayed by the phase shifters 807a2, 807a3, and 807a4, by a phase shift amount 3Φ.
In other words, a direction of the maximum sensitivity for radio waves received by the antenna elements 806a-806d is a direction D that forms a predetermined angle Θ(Θ=cos−1(Φ/d)) with respect to the direction of the row of the antenna elements 806a-806d. It is assumed here that reference numerals w1 to w3 in FIG. 10(a) denote planes of the received waves in the same phase, respectively.
However, in the conventional phased-array antenna 803 having the above-mentioned construction, the numbers of phase shifters 807 which are located between the respective antenna elements 806 and the input terminal 808 are different, and further, there are transmission losses in the respective phase shifters 807. Therefore, the effects of combining powers from the respective antenna elements 806a-806d are decreased, so that the shape of the beam that is shown in FIG. 10(b) is deformed, whereby it is difficult to obtain a pointed beam (large directivity gain). In addition, the amount of beam tilt is reduced, and as a result, the control of the antenna's directivity is deteriorated.
Further, as described with reference to FIG. 9(a), each of the phase shifters 807 that are used for the conventional phased-array antenna 830 is formed in one piece, by allocating areas on the same plane to the ferroelectric base material 702 and the paraelectric base material 701 which constitute the phase shifter 700, respectively. Therefore, a distributed capacitance Cn per unit length of the line for the microstrip hybrid coupler 703 and a distributed capacitance Cf per unit length of the line for the microstrip stub 704 are greatly different from each other. Accordingly, high-frequency power reflection is produced at the connection between the microstrip hybrid coupler 703 and the microstrip stub 704, whereby the power from the microstrip hybrid coupler 703 does not enter the microstrip stub 704 so efficiently, and consequently, the sufficient phase shift amount cannot be obtained.
Hereinafter, a detailed explanation will be given. For, example, the line impedance Z is generally expressed by the distributed inductance L per unit length of the line and the distributed capacitance C per unit length of the line as Z^2 (the square of Z)=L/C. Further, when it is assumed that all fields exist only within the base material, and all of the fields are approximated to be linear and perpendicular to the ground conductor, the distributed capacitance C per unit length of the line is expressed by the line width W, the base material thickness H, and the base material permittivity ε, as C=εW/H. When the distributed capacitance Cn per unit length of the line for the microstrip hybrid coupler 703 and the distributed capacitance Cf per unit length of the line for the microstrip stub 704 are compared with each other by utilizing the above-mentioned expressions, assuming that the permittivity of the paraelectric base material 701 as the base material of the microstrip hybrid coupler 703 is εn and the permittivity of the ferroelectric base material 702 as the base material of the microstrip stub 704 is εf, the relationship εn<<εf is generally established. Further, since the line widths W of the microstrip hybrid coupler 703 and the microstrip stub 704, and the distances H of the respective conductors are the same, the distributed capacitance Cn per unit length of the line for the microstrip hybrid coupler 703 (=εnW/H) and the distributed capacitance Cf per unit length of the line for the microstrip stub 704 (=εfW/H) are greatly different. Consequently, as mentioned above, the power from the microstrip hybrid coupler 703 does not enter the microstrip stub 704 so efficiently, and thus, the sufficient phase shift amount cannot be obtained.
To overcome this problem, the method in which a magnetic material is provided in proximity of the microstrip stub 704 to increase the distributed inductance L per unit length of the line for the microstrip stub 704, thereby enhancing the line impedance Z, is disclosed in the above-mentioned Prior Art 1, and its construction is also suggested therein.
However, when the magnetic material is provided in proximity of the microstrip stub 704 of the phase shifter 700 to suppress the reduction in the matching degree of the line impedance Z between both the line sections 703 and 704, so as to obtain a larger phase shift amount, as in the above-mentioned Prior Art 1, there arises an additional problem in that more processes are needed when the phase shifter 700 is produced by firing. As a result, the manufacturing cost of the phase shifter is adversely increased.
The present invention is made to solve the above-mentioned problems. Accordingly, an object of the present invention is to provide an antenna control unit that can be manufactured in fewer manufacturing processes (low cost), and has a pointed beam (large directivity gain) and a large amount of beam tilt, and a phased-array antenna that employs such an antenna control unit.